EMC FLEX BLOG A site dedicated to Automotive EMC Testing for Electronic Modules

Differential Mode Current vs Common Mode Current (Transmission Lines)

15. December 2020 12:14 by Christian in EMC/EMI, Noise Coupling, Troubleshooting
Differential Mode Configuration Assuming 1A is propagated from the source to the load usin

See Ground Return & Common Impedance Coupling

Differential Mode Configuration

Assuming 1A is propagated from the source to the load using I1 to represent the current flow. The 1A current must return to the source represented by I2. If I1 = I2 then we have a perfectly balanced transmission line system, no loss in the network.
The EM filed that exists in the outgoing path will couple inductively to the RF return path (AC transmission while DC will always travel in the lowest rsistance path I2). Magnetic flux between these two transmission lines will cancel each other out, being of equal value and opposite in dirrection. Assuming that the spacing between opposite conductors is very small, there should be no radiated emissions. Differential-mode radiation is caused by the flow of RF current loops within a system 's structure.
Common Mode Configuration
Assuming tht 50% of the transmitted current is consumed within the load, it leaves 50% of current that must be returned to its source.  The Kirchhoff's Law states that the sum of all currents withinn a transmission line must equal zero.We have 50% loss. 
I'2 represents the a virtual return path through free space or metallic interconnect. Not all desired return current will flow in I2 due to inductance or loss in transmission line. The remaining of the desired return current will flow in I'2. A negative current flow will exist in I2, travelling in opposite direction to satisfy Ampere's Law. The undesired (negative) current flow in I2 is that portion that contributes to common-mode currents.
Common mode radiation results from unintentional voltage drops caused by a circuit rising above the 0V reference.
Cables connected to the affected reference system will act as dipole antenna when stimulated with a voltage source.
The only solution to resolve CM radiation is reducing the common path impedance for  the return current.
The total magnitude of imbalance in a DM transmission line system becomes the the total magnitude of CM current.
RF loss within a system or transmission line will result in CM energy, and this CM current is the reason for EMI problems.

Grounding for Automotive EMC Load Simulators

15. December 2020 09:02 by Christian in EMC/EMI, EMC TEST PLAN, Grounding
The Load Simulator must be robust and as simple as possible to become a valid reference for DUT EMC

The Load Simulator must be robust and as simple as possible to become a valid reference for DUT EMC performace evaluation. The most common mistake during LS configuration for RE, BCI, RI ALSE is related to how DUT's supply return is interconected with the rest of DUT support equipment. Incorrect grounding between DUT, Load Simulator, Support Equipment, Ground Plane, dedicated Earth Grounding Rod, and Buildin Safety Ground can end up in unwanted grounding loops or as shown below to a situation where the GND LISN Input is connected to GND LISN Output.

An ideal Load Simulator is just a pass-through enclosure with test points, control switches, no active electronics.  Most of the time the DUT is powered straight from the output of the B+ LISN  or a Pulse Generator following certain rules in terms of B+ and GND leads length. The input of the LISN for battery negative pole is always connected to ground plane. Depending on the OEM specification or international standard used, the Load Simulator is powered directly from the automotive battery or from the output of the B+ LISN. If powered from the output of the LISN, the active electronic components part of the LS can play a role in the EMC compliance of the DUT.  In automotive EMC each test bench or EMC test chamber should have dedicated Eart Grounding Rod completely separated from the Buliding Safety Ground. The incorrect grounding configuration below shows how via the test ground plane the building safety ground is in contact with the dedicated earth grounding rod. In this situation the output of the LISN is shorted to its input cancelling the purpose of the LISN.

Never connect the negative terminal from support equipment power supplies to their terminal for safety ground. 

 2020-12-14 Christian Rosu


PCB Signal Return & Power Return Planes

14. December 2020 11:39 by Christian in EMC/EMI, Grounding, PCB
Diverting a return current path over a longer route can cause both radiated emissions and

See Differential Mode vs Common Mode Current

Diverting a return current path over a longer route can cause both radiated emissions and RF immunity issues. At frequencies above 100 kHz, the return current flows along the path of least impedance (e.g. directly under the signal or clock trace).

Splitting the analog and digital return planes, noisy digital return currents will stay out of the sensitive analog area. Runing digital signal traces across isolated analog areas can contaminate the analog area. The two planes are generally connected together at the PC board power connector.

In a scenario with a single return plane the critical part is routing the signal traces (and corresponding return currents) so they don't cross the A/D boundary. 

Stitching capacitors allow a path for return currents to get back to the source when crossing multiple planes with differing potentials (e.g. power and signal return planes). They need to be located as closely as possible to where the high frequency trace penetrates the planes. The value is not critical (1 to 10 nF), but should present a low impedance at the frequency in question (plus harmonics). 

Multiple vias will provide multiple paths back to the source. At really high frequencies (above 500 MHz into the GHz region), the power and power return planes can form a cavity resonance and causeradiated emissions. Adding a pattern of stitching capacitors can help break up this resonance. There are also experiments on the use of “lossy” bypass capacitors (high ESR) mounted around the board that serve to damp the resonances. 

Approaching frequencies above 100 MHz, the series inductance can become significant. Therefore a classic via would work better than a zero-Ohm resistor, depending on the connecting traces. 

Number of PCB Layers

From an EMC standpoint, eight, or more, layers has proven best. The problem with four or six-layer board designs is that it becomes very difficult to define a solid lowimpedance return path when running high speed signals and clock traces through multiple power/return planes. The power and signal/power return planes to be as close together as possible and sometimes this is difficult to

Trace Length

The general rule of thumb is that if a trace (or cable) is electrically 1/20th wavelength, or less, then it becomes a very inefficient radiating structure. As the length starts to approach a half-wavelength, then it becomes an efficient antenna. 

Capacitive Coupling vs Inductive Coupling

13. December 2020 19:29 by Christian in EMC/EMI, Noise Coupling, Troubleshooting
Electromagnetic fields and how they propagate and couple together. Conductive coupling is usually lo

How Electromagnetic Fields are propagated and coupled together.

Conductive coupling is usually low frequency noise traveling along a pair of wires due to a common resistive path, which form a loop. Breaking the connection stops the interference (“ground loops”).

Radiated coupling represent EM waves propagating through space from one point to another, whose coupling factor reduces by a factor of 1/r.

Inductive and Capacitive couplig refers to near field effects, which are reduced rapidly with separation (near field terms of 1/r^2 and 1/r^3). 

Inductive coupling requires two "loops" coupled together, with the field of the one inducing noise into the other (think transformer).

Example: one cable inducing noise into an adjacent cable.

Inductive coupling is a high di/dt phenomenon.

Capacitive coupling requires two “plates” with the noisy plate inducing a voltage change in the other due to dispersion effects (think capacitor).

Example: heat sink from a switched mode power supply inducing voltage changes in a nearby cable or chassis.

Capacitive coupling is generally a high dv/dt phenomenon.

One-Meter Horizontal Distance Antenna to DUT Radiated Emissions Measurements

The horizontal distance between DUT and Antenna for automotive EMC compliance is 1 meter. For other

The horizontal distance between DUT and Antenna for automotive EMC compliance is 1 meter. For other non-automotive regulatory and standard-based measurements these distances are 3m, 10m, or 30m.

One-meter DUT-to-antenna distance measurements are carried out based on MIL-STD 461 (military), RTCA DO-160 & EUROCAE ED-14 (commercial aircraft), and CISPR 25 (automotive).

CISPR 25 specifies a one-meter antenna distance to be used for radiated emissions from Components/Modules in an Absorber Lined Shielded Enclosure (ALSE).


The near field and far field are regions of the electromagnetic field (EM) around an object.
Far-field E (electric) and B (magnetic) field strength decreases as the distance from the source increases, resulting in an inverse-square law for the radiated power intensity of electromagnetic radiation.

Near-field E (electric) and B (magnetic) field strength decrease more rapidly with distance:

  • the radiative field decreases by the inverse-distance squared, resulting in a diminished power in the parts of the electric field by an inverse fourth-power
  • the reactive field by an inverse cubed law, resulting in a diminished power in the parts of the electric field by an inverse sixth-power

The rapid drop in power contained in the near-field ensures that effects due to the near-field essentially vanish a few wavelengths away from the radiating part of the antenna.

dF = (2* D^2)/λ
D= largest dimension of the radiator or diameter of antenna
λ = wavelength of the radio wave
dF = 2*(D/λ)^2
λ = 2* (D/λ)*D
dF >> D
dF >> λ

Near-field and far-field regions for an antenna (diameter or length D) larger than  the wavelength of the radiation it emits, so that ​D⁄λ ≫ 1:

  • Near Field
    R = near field antenna to radiating filed distance
    R = 0.62 * (D^3/λ)^1/2
  • Far Field
    Ro = far field antenna to radiating filed distance
    Ro = 2*(D^2/ λ)